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  • 型号: AD7569JRZ
  • 制造商: Analog
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AD7569JRZ产品简介:

ICGOO电子元器件商城为您提供AD7569JRZ由Analog设计生产,在icgoo商城现货销售,并且可以通过原厂、代理商等渠道进行代购。 AD7569JRZ价格参考。AnalogAD7569JRZ封装/规格:接口 - 专用, Analog I/O Interface 24-SOIC。您可以下载AD7569JRZ参考资料、Datasheet数据手册功能说明书,资料中有AD7569JRZ 详细功能的应用电路图电压和使用方法及教程。

产品参数 图文手册 常见问题
参数 数值
产品目录

集成电路 (IC)半导体

描述

IC I/O PORT 8BIT ANALOG 24-SOIC模数转换器 - ADC 8-BIT CMOS I/O PORT w/ SGL Output

产品分类

接口 - 专用

品牌

Analog Devices

产品手册

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产品图片

rohs

符合RoHS无铅 / 符合限制有害物质指令(RoHS)规范要求

产品系列

数据转换器IC,模数转换器 - ADC,Analog Devices AD7569JRZLC²MOS

数据手册

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产品型号

AD7569JRZ

产品目录页面

点击此处下载产品Datasheet

产品种类

模数转换器 - ADC

供应商器件封装

24-SOIC W

信噪比

44 dB

分辨率

8 bit

包装

管件

商标

Analog Devices

安装类型

表面贴装

安装风格

SMD/SMT

封装

Tube

封装/外壳

24-SOIC(0.295",7.50mm 宽)

封装/箱体

SOIC-24

工作电源电压

5 V

工厂包装数量

31

应用

模拟 I/O

接口

总线

接口类型

Parallel

最大功率耗散

85 mW

最大工作温度

+ 125 C

最小工作温度

- 55 C

标准包装

31

电压-电源

4.75 V ~ 5.25 V

电压参考

Internal

系列

AD7569

结构

SAR

转换器数量

1

转换速率

500 kS/s

输入类型

Single-Ended

通道数量

1 Channel

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PDF Datasheet 数据手册内容提取

a LC2MOS Complete, 8-Bit Analog I/0 Systems AD7569/AD7669 FEATURES AD7569 FUNCTIONAL BLOCK DIAGRAM 2 (cid:109)s ADC with Track/Hold 1 (cid:109)s DAC with Output Amplifier AD7569, Single DAC Output AD7669, Dual DAC Output On-Chip Bandgap Reference Fast Bus Interface Single or Dual 5 V Supplies GENERAL DESCRIPTION The AD7569/AD7669 is a complete, 8-bit, analog I/O system on a single monolithic chip. The AD7569 contains a high speed successive approximation ADC with 2 m s conversion time, a track/ AD7669 FUNCTIONAL BLOCK DIAGRAM hold with 200 kHz bandwidth, a DAC and an output buffer ampli- fier with 1 m s settling time. A temperature-compensated 1.25 V bandgap reference provides a precision reference voltage for the ADC and the DAC. The AD7669 is similar, but contains two DACs with output buffer amplifiers. A choice of analog input/output ranges is available. Using a sup- ply voltage of +5 V, input and output ranges of zero to 1.25 V and zero to 2.5 volts may be programmed using the RANGE in- put pin. Using a – 5 V supply, bipolar ranges of – 1.25 V or – 2.5 V may be programmed. Digital interfacing is via an 8-bit I/O port and standard micro- processor control lines. Bus interface timing is extremely fast, al- lowing easy connection to all popular 8-bit microprocessors. A separate start convert line controls the track/hold and ADC to give precise control of the sampling period. PRODUCT HIGHLIGHTS The AD7569/AD7669 is fabricated in Linear-Compatible 1. Complete Analog I/O on a Single Chip. CMOS (LC2MOS), an advanced, mixed technology process The AD7569/AD7669 provides everything necessary to combining precision bipolar circuits with low power CMOS interface a microprocessor to the analog world. No external logic. The AD7569 is packaged in a 24-pin, 0.3" wide “skinny” components or user trims are required and the overall accu- DIP, a 24-terminal SOIC and 28-terminal PLCC and LCCC racy of the system is tightly specified, eliminating the need packages. The AD7669 is available in a 28-pin, 0.6" plastic to calculate error budgets from individual component DIP, 28-terminal SOIC and 28-terminal PLCC package. specifications. 2. Dynamic Specifications for DSP Users. In addition to the traditional ADC and DAC specifications, the AD7569/AD7669 is specified for ac parameters, includ- ing signal-to-noise ratio, distortion and input bandwidth. 3. Fast Microprocessor Interface. The AD7569/AD7669 has bus interface timing compatible with all modern microprocessors, with bus access and relin- quish times less than 75 ns and write pulse width less than 80 ns. REV.B Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. which may result from its use. No license is granted by implication or Tel: 617/329-4700 World Wide Web Site: http://www.analog.com otherwise under any patent or patent rights of Analog Devices. Fax: 617/326-8703 © Analog Devices, Inc., 1996

AD7569/AD7669–SPECIFICATIONS DAC SPECIFICATIONS1 (VDD = +5 V (cid:54) 5%; VSS2 = RANGE = AGNDDAC = AGNDADC = DGND = 0 V; RL = 2 k(cid:86), CL = 100 pF to AGNDDAC unless otherwise noted. All specifications T to T unless otherwise noted.) MIN MAX AD7569 J, A Versions3 AD7569 AD7669 K, B AD7569 AD7569 Parameter J Version Versions S Version T Version Units Conditions/Comments STATIC PERFORMANCE Resolution4 8 8 8 8 Bits Total Unadjusted Error5 – 2 – 2 – 3 – 3 LSB typ Relative Accuracy5 – 1 – 1/2 – 1 – 1/2 LSB max Differential Nonlinearity5 – 1 – 3/4 – 1 – 3/4 LSB max Guaranteed Monotonic Unipolar Offset Error DAC data is all 0s; VSS = 0 V @ +25(cid:176)C – 2 – 1.5 – 2 – 1.5 LSB max Typical tempco is 10 m V/(cid:176)C for +1.25 V range TMIN to TMAX – 2.5 – 2 – 2.5 – 2 LSB max Bipolar Zero Offset Error DAC data is all 0s; VSS = –5 V @ +25(cid:176)C – 2 – 1 5 – 2 – 1.5 LSB max Typical tempco is 20 m V/(cid:176)C for – 1.25 V range TMIN to TMAX – 2.5 – 2 – 2.5 – 2 LSB max Full-Scale Error6 (AD7569 Only) VDD = 5 V @ +25(cid:176)C – 2 – 1 – 2 – 1 LSB max TMIN to TMAX – 3 – 2 – 4 – 3 LSB max Full-Scale Error6 (AD7669 Only) VDD = 5 V @ +25(cid:176)C – 3 LSB max TMIN to TMAX – 4.5 LSB max DACA/DACB Full-Scale Error Match6 (AD7669 Only) – 2.5 LSB max VDD = 5 V D Full Scale/D VDD, TA = +25(cid:176)C 0.5 0.5 0.5 0.5 LSB max VOUT = 2.5 V; D VDD = – 5% D Full Scale/D VSS, TA = +25(cid:176)C 0.5 0.5 0.5 0.5 LSB max VOUT = –2.5 V; D VSS = – 5% Load Regulation at Full Scale 0.2 0.2 0.2 0.2 LSB max RL = 2 kW to (cid:176)/C DYNAMIC PERFORMANCE Signal-to-Noise Ratio5 (SNR) 44 46 44 46 dB min VOUT = 20 kHz full-scale sine wave with fSAMPLING = 400 kHz Total Harmonic Distortion5 (THD) 48 48 48 48 dB max VOUT = 20 kHz full-scale sine wave with fSAMPLING = 400 kHz Intermodulation Distortion5 (IMD) 55 55 55 55 dB typ fa = 18.4 kHz, fb = 14.5 kHz with fSAMPLING = 400 kHz ANALOG OUTPUT Output Voltage Ranges Unipolar 0 to +1.25/2.5 Volts VDD = +5 V, VSS = 0 V Bipolar – 1.25/– 2.5 Volts VDD = +5 V, VSS = –5 V LOGIC INPUTS CS, X/B,WR, RANGE, RESET, DB0–DB7 Input Low Voltage, VINL 0.8 0.8 0.8 0.8 V max Input High Voltage, VINH 2.4 2.4 2.4 2.4 V min Input Leakage Current 10 10 10 10 m A max VIN = 0 to VDD Input Capacitance7 10 10 10 10 pF max DB0–DB7 Input Coding (Single Supply) Binary Input Coding (Dual Supply) 2s Complement AC CHARACTERlSTICS7 Voltage Output Settling Time Settling time to within – 1/2 LSB of final value Positive Full-Scale Change 2 2 2 2 m s max Typically 1 m s Negative Full-Scale Change (Single Supply) 4 4 4 4 m s max Typically 2 m s Negative Full-Scale Change (Dual Supply) 2 2 2 2 m s max Typically 1 m s Digital-to-Analog Glitch Impulse5 15 15 15 15 nV secs typ Digital Feedthrough5 1 1 1 1 nV secs typ VIN to VOUT Isolation 60 60 60 60 dB typ VIN = – 2.5 V, 50 kHz Sine Wave DAC to DAC Crosstalk5 (AD7669 Only) 1 nV secs typ DACA to DACB Isolation5 (AD7669 Only) –70 dB max POWER REQUIREMENTS VDD Range 4.75/5.25 4.75/5.25 4.75/5.25 4.75/5.25 V min/V max For Specified Performance VSS Range (Dual Supplies) –4.75/–5.25 –4.75/–5.25 –4.75/–5.25 –4.75/–5.25 V min/V max Specified Performance also applies to VSS = 0 V for unipolar ranges. IDD VOUT = VIN = 2.5 V; Logic Inputs = 2.4 V; CLK = 0.8 V (AD7569) 13 13 13 13 mA max Output unloaded (AD7669) 18 mA max Outputs unloaded ISS (Dual Supplies) VOUT = VIN = –2.5 V; Logic Inputs = 2.4 V; CLK = 0.8 V (AD7569) 4 4 4 4 mA max Output unloaded (AD7669) 6 mA max Outputs unloaded DAC/ADC MATCHING Gain Matching6 VIN to VOUT match with VIN = – 2.5 V, @ +25(cid:176)C 1 1 1 1 % typ 20 kHz sine wave TMIN to TMAX 1 1 1 1 % typ NOTES 1Specifications apply to both DACs in the AD7669. VOUT applies to both VOUTA and VOUTB of the AD7669. 2Except where noted, specifications apply for all output ranges including bipolar ranges with dual supply operation. 3Temperature ranges as follows: J, K versions; 0(cid:176)C to +70(cid:176)C A, B versions; –40(cid:176)C to +85(cid:176)C S, T versions; –55(cid:176)C to +125(cid:176)C 41 LSB = 4.88 mV for 0 V to +1.25 V output range, 9.76 mV for 0 V to +2.5 V and – 1.25 V ranges and 19.5 mV for – 2.5 V range. 5See Terminology. 6Includes internal voltage reference error and is calculated after offset error has been adjusted out. Ideal unipolar full-scale voltage is (FS – 1 LSB); ideal bipolar positive full-scale voltage is (FS/2 – 1 LSB) and ideal bipolar negative full-scale voltage is –FS/2. 7Sample tested at +25(cid:176)C to ensure compliance. Specifications subject to change without notice. –2– REV. B

AD7569/AD7669 ADC SPECIFICATIONS (V = +5 V (cid:54) 5%; V 1 = RANGE = AGND = AGND = DGND = 0 V; f = 5 MHz external unless other- DD SS DAC DAC CLK wise noted. All specifications T to T unless otherwise noted.) Specifications apply to Mode 1 interface. MIN MAX AD7569 J, A Versions3 AD7569 AD7669 K, B AD7569 AD7569 Parameter J Version Versions S Version T Version Units Conditions/Comments DC ACCURACY Resolution3 8 8 8 8 Bits Total Unadjusted Error4 – 3 – 3 – 4 – 4 LSB typ Relative Accuracy4 – 1 – 1/2 – 1 – 1/2 LSB max Differential Nonlinearity4 – 1 – 3/4 – 1 – 3/4 LSB max No Missing Codes Unipolar Offset Error Typical tempco is 10 m V/(cid:176)C for +1.25 V range; VSS = 0 V @ +25(cid:176)C – 2 – 1.5 – 2 – 1.5 LSB max TMIN to TMAX – 3 – 2.5 – 3 – 2.5 LSB max Bipolar Zero Offset Error Typical tempco is 20 m V/(cid:176)C for + 1.25 V range; VSS = –5 V @ +25(cid:176)C – 3 – 2.5 – 3 – 2.5 LSB max TMIN to TMAX – 3.5 – 3 – 4 – 3.5 LSB max Full-Scale Error5 VDD = 5 V @ +25(cid:176)C –4, +0 –4, +0 –4, +0 –4, +0 LSB max TMIN to TMAX –5.5, +1.5 –5.5, +1.5 –7.5, +2 –7.5, +2 LSB max D Full Scale/D VDD, TA = +25(cid:176)C 0.5 0.5 0.5 0.5 LSB max VIN = +2.5 V; D VDD = – 5% D Full Scale/D VSS, TA = +25(cid:176)C 0.5 0.5 0.5 0.5 LSB max VIN = –2.5 V; D VSS = – 5% DYNAMIC PERFORMANCE Signal-to-Noise Ratio4 (SNR) 44 46 44 45 dB min VIN = 100 kHz full-scale sine wave with fSAMPLING = 400 kHz6 Total Harmonic Distortion4 (THD) 48 48 48 48 dB max VIN = 100 kHz full-scale sine wave with fSAMPLING = 400 kHz6 Intermodulation Distortion4 (IMD) 60 60 60 60 dB typ fa = 99 kHz, fb = 96.7 kHz with fSAMPLING = 400 kHz Frequency Response 0.1 0.1 0.1 0.1 dB typ VIN = – 2.5 V, dc to 200 kHz sine wave Track/Hold Acquisition Time7 200 200 300 300 ns typ ANALOG INPUT Input Voltage Ranges Unipolar 0 to +1.25/ +2.5 Volts VDD = +5 V; VSS = 0 V Bipolar – 1.25/– 2.5 Volts VDD = +5 V; VSS = –5 V Input Current – 300 – 300 – 300 – 300 m A max See equivalent circuit Figure 5 Input Capacitance 10 10 10 10 pF typ LOGIC INPUTS CS, RD, ST, CLK, RESET, RANGE Input Low Voltage, VINL 0.8 0.8 0.8 0.8 V max Input High Voltage, VINH 2.4 2.4 2.4 2.4 V min Input Capacitance8 10 10 10 10 pF max CS, RD, ST, RANGE, RESET Input Leakage Current 10 10 10 10 m A max V IN = 0 to VDD CLK Input Current IINL –1.6 –1.6 –1.6 –1.6 mA max VIN = 0 V IINH 40 40 40 40 m A max VIN = VDD LOGIC OUTPUTS DB0–DB7, INT, BUSY VOL, Output Low Voltage 0.4 0.4 0.4 0.4 V max ISINK = 1.6 mA VOH, Output High Voltage 4.0 4.0 4.0 4.0 V min ISOURCE = 200 m A DB0–DB7 Floating State Leakage Current 10 10 10 10 m A max Floating State Output Capacitance8 10 10 10 10 pF max Output Coding (Single Supply) Binary Output Coding (Dual Supply) 2s Complement CONVERSION TIME With External Clock 2 2 2 2 m s max fCLK = 5 MHz With Internal Clock, TA = +25(cid:176)C 1.6 1.6 1.6 1.6 m s min Using recommended clock components shown in Figure 21. 2.6 2.6 2.6 2.6 m s max Clock frequency can be adjusted by varying RCLK. POWER REQUIREMENTS As per DAC Specifications NOTES 1Except where noted, specifications apply for all ranges including bipolar ranges with dual supply operation. 2Temperature ranges are as follows:J, K versions; 0(cid:176)C to +70(cid:176)C A, B versions; –40(cid:176)C to +85(cid:176)C S, T versions; –55(cid:176)C to +125(cid:176)C 31 LSB = 4.88 mV for 0 V to +1.25 V range, 9.76 mV for 0 V to +2.5 V and – 1.25 V ranges and 19.5 mV for +2.5 V range. 4See Terminology. 5Includes internal voltage reference error and is calculated after offset error has been adjusted out. Ideal unipolar last code transition occurs at (FS – 3/2 LSB). Ideal bipolar last code transition occurs at (FS/2 – 3/2 LSB). 6Exact frequencies are 101 kHz and 384 kHz to avoid harmonics coinciding with sampling frequency. 7Rising edge of BUSY to falling edge of ST. The time given refers to the acquisition time, which gives a 3 dB degradation in SNR from the tested figure. 8Sample tested at +25(cid:176)C to ensure compliance. Specifications subject to change without notice. REV. B –3–

AD7569/AD7669–TIMING CHARACTERISTICS1 (See Figures 8, 10, 12; V = 5 V (cid:54) 5%; V = 0 V or –5 V (cid:54) 5%) DD SS Limit at Limit at Limit at T , T T , T MIN MAX MIN MAX Parameter 25(cid:56)C (All Grades) (J, K, A, B Grades) (S, T Grades) Units Test Conditions/Comments DAC Timing t 80 80 90 ns min WR Pulse Width 1 t 0 0 0 ns min CS, A/B to WR Setup Time 2 t 0 0 0 ns min CS, A/B to WR Hold Time 3 t 60 70 80 ns min Data Valid to WR Setup Time 4 t 10 10 10 ns min Data Valid to WR Hold Time 5 ADC Timing t 50 50 50 ns min ST Pulse Width 6 t 110 130 150 ns max ST to BUSY Delay 7 t 20 30 30 ns max BUSY to INT Delay 8 t 0 0 0 ns min BUSY to CS Delay 9 t 0 0 0 ns min CS to RD Setup Time 10 t 60 75 90 ns min RD Pulse Width Determined by t . 11 13 t 0 0 0 ns min CS to RD Hold Time 12 t 2 60 75 90 ns max Data Access Time after RD; C = 20 pF 13 L 95 120 135 ns max Data Access Time after RD; C = 100 pF L t 3 10 10 10 ns min Bus Relinquish Time after RD 14 60 75 85 ns max t 65 75 85 ns max RD to INT Delay 15 t 120 140 160 ns max RD to BUSY Delay 16 t 2 60 75 90 ns max Data Valid Time after BUSY; C = 20 pF 17 L 90 115 135 ns max Data Valid Time after BUSY; C = 100 pF L NOTES 1Sample tested at +25(cid:176)C to ensure compliance. All input control signals are specified with t = t = 5 ns (10% to 90% of +5 V) and timed from a voltage level of 1.6 V. R F 2t and t are measured with the load circuits of Figure 1 and defined as the time required for an output to cross either 0.8 V or 2.4 V. 13 17 3t is defined as the time required for the data line to change 0.5 V when loaded with the circuit of Figure 2. l4 Specifications subject to change without notice. a. High-Z to V b. High-Z to V a. V to High-Z b. V to High-Z OH OL OH OL Figure 1.Load Circuits for Data Access Time Test Figure 2.Load Circuits for Bus Relinquish Time Test ABSOLUTE MAXIMUM RATINGS V to AGND or AGND . . . . . . . . . . . . .–0.3 V, +7 V Power Dissipation (Any Package) to +75(cid:176) C . . . . . . . .450 mW VDD to DGNDD .A C. . . . . . . . . .A .D .C . . . . . . . . . . . . . .–0.3 V, +7 V Derates above 75(cid:176) C by . . . . . . . . . . . . . . . . . . . . . 6 mW/(cid:176) C DD Operating Temperature Range V to V . . . . . . . . . . . . . . . . . . . . . . . . . . . . .–0.3 V, +14 V DD SS Commercial (J, K) . . . . . . . . . . . . . . . . . . . . . .0(cid:176) C to +70(cid:176) C AGND or AGND to DGND . . . .–0.3 V, V + 0.3 V AGNDDAC to AGNDADC . . . . . . . . . . . . . . . . . . . . .D .D . . .– 5 V Industrial (A, B) . . . . . . . . . . . . . . . . . . . . . –40(cid:176) C to +85(cid:176) C DAC ADC Extended (S, T) . . . . . . . . . . . . . . . . . . . . –55(cid:176) C to +125(cid:176) C Logic Voltage to DGND . . . . . . . . . . . . .–0.3 V, V + 0.3 V DD Storage Temperature Range . . . . . . . . . . . . –65(cid:176) C to +150(cid:176) C CLK Input Voltage to DGND . . . . . . . . .–0.3 V, V + 0.3 V DD Lead Temperature (Soldering, 10 secs) . . . . . . . . . . . +300(cid:176) C V (V A, V B) to OUT OUT OUT AGND1 . . . . . . . . . . . . . . . . . V – 0.3 V, V + 0.3 V *Stresses above those listed under “Absolute Maximum Ratings” may cause DAC SS DD V to AGND . . . . . . . . . . . . . . . V – 0.3 V, V + 0.3 V permanent damage to the device. This is a stress rating only; functional operation IN ADC SS DD of the device at these or any other condition above those indicated in the NOTE operational sections of this specification is not implied. Exposure to absolute 1Output may be shorted to any voltage in the range V to V provided that the SS DD maximum rating conditions for extended periods may affect device reliability. power dissipation of the package is not exceeded. Typical short circuit current for a short to AGND or V is 50 mA. SS CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000V readily WARNING! accumulate on the human body and test equipment and can discharge without detection. Although the AD7569/AD7669 features proprietary ESD protection circuitry, permanent dam- age may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. ESD SENSITIVE DEVICE –4– REV. B

AD7569/AD7669 NOTE: Digital Feedthrough The term DAC (Digital-to-Analog Converter) throughout the Digital Feedthrough is also a measure of the impulse injected to data sheet applies equally to the dual DACs in the AD7669 as the analog output from the digital inputs, but is measured when well as to the single DAC of the AD7569 unless otherwise the DAC is not selected. It is essentially feedthrough across the stated. It follows that the term V applies to both V A and die and package. It is also a measure of the glitch impulse trans- OUT OUT V B of the AD7669 also. ferred to the analog output when data is read from the internal OUT ADC. It is specified in nV secs and is measured with WR high TERMINOLOGY and a digital code change from all 0s to all 1s. Total Unadjusted Error Total unadjusted error is a comprehensive specification that in- DAC-to-DAC Crosstalk (AD7669 Only) cludes internal voltage reference error, relative accuracy, gain The glitch energy transferred to the output of one DAC due to and offset errors. an update at the output of the second DAC. The figure given is the worst case and is expressed in nV secs. It is measured with Relative Accuracy (DAC) an update voltage of full scale. Relative Accuracy or endpoint nonlinearity is a measure of the maximum deviation from a straight line passing through the DAC-to-DAC Isolation (AD7669 Only) endpoints of the DAC transfer function. It is measured after al- DAC-to-DAC Isolation is the proportion of a digitized sine lowing for offset and gain errors. For the bipolar output ranges, wave from the output of one DAC, which appears at the output the endpoints of the DAC transfer function are defined as those of the second DAC (loaded with all 1s). The figure given is the voltages that correspond to negative full-scale and positive full- worst case for the second DAC output and is expressed as a ra- scale codes. For the unipolar output ranges, the endpoints are tio in dBs. It is measured with a digitized sine wave (f = SAMPLING code 1 and code 255. Code 1 is chosen because the amplifier is 100 kHz) of 20 kHz at 2.5 V pk-pk. now working in single supply and, in cases where the true offset Signal-to-Noise Ratio of the amplifier is negative, it cannot be seen at code 0. If the Signal-to-Noise Ratio (SNR) is the measured signal to noise at relative accuracy were calculated between code 0 and code 255, the output of the converter. The signal is the rms magnitude of the “negative offset” would appear as a linearity error. If the off- the fundamental. Noise is the rms sum of all the nonfundamen- set is negative and less than 1 LSB, it will appear at code 1, and tal signals (excluding dc) up to half the sampling frequency. hence the true linearity of the converter is seen between code 1 SNR is dependent on the number of quantization levels used in and code 255. the digitization process; the more levels, the smaller the quanti- Relative Accuracy (ADC) zation noise. The theoretical SNR for a sine wave is given by Relative Accuracy is the deviation of the ADC’s actual code SNR = (6.02N + 1.76) dB transition points from a straight line drawn between the end- where N is the number of bits. Thus for an ideal 8-bit converter, points of the ADC transfer function. For the bipolar input SNR = 50 dB. ranges, these points are the measured, negative, full-scale transi- Harmonic Distortion tion point and the measured, positive, full-scale transition point. Harmonic Distortion is the ratio of the rms sum of harmonics to For the unipolar ranges, the straight line is drawn between the the fundamental. For the AD7569/AD7669, Total Harmonic measured first LSB transition point and the measured full-scale Distortion (THD) is defined as transition point. Differential Nonlinearity (cid:86) (cid:50)+(cid:86) (cid:50)+(cid:86) (cid:50)+(cid:86) (cid:50)+(cid:86) (cid:50) Differential Nonlinearity is the difference between the measured (cid:50)(cid:48)(cid:108)(cid:111)(cid:103) (cid:50) (cid:51) (cid:52) (cid:53) (cid:54) (cid:86) change and an ideal 1 LSB change between any two adjacent (cid:49) codes. A specified differential nonlinearity of – 1 LSB max en- where V1 is the rms amplitude of the fundamental and V2, V3, V , V and V are the rms amplitudes of the individual sures monotonicity (DAC) or no missed codes (ADC). A differ- 4 5 6 ential nonlinearity of – 3/4 LSB max ensures that the minimum harmonics. step size (DAC) or code width (ADC) is 1/4 LSB, and the maxi- Intermodulation Distortion mum step size or code width is 3/4 LSB. With inputs consisting of sine waves at two frequencies, fa and fb, any active device with nonlinearities will create distortion Digital-to-Analog Glitch Impulse products, of order (m + n), at sum and difference frequencies of Digital-to-Analog Glitch Impulse is the impulse injected into the mfa – nfb where m, n = 0, l, 2, 3,… . Intermodulation terms analog output when the digital inputs change state with the are those for which m or n is not equal to zero. For example, DAC selected. It is normally specified as the area of the glitch in the second order terms include (fa + fb) and (fa – fb) and the nV secs and is measured when the digital input code is changed third order terms include (2fa + fb), (2fa – fb), (fa + 2fb) and by 1 LSB at the major carry transition. (fa – 2fb). REV. B –5–

AD7569/AD7669 AD7569 PIN CONFIGURATIONS DIP, SOIC PLCC LCCC AD7669 PIN CONFIGURATIONS DIP, SOIC ORDERING GUIDE Relative Temperature Accuracy Package Model Range T –T Option1 MIN MAX AD7569JN 0(cid:176) C to +70(cid:176) C – 1 LSB N-24 AD7569JR 0(cid:176) C to +70(cid:176) C – 1 LSB R-24 AD7569AQ –40(cid:176) C to +85(cid:176) C – 1 LSB Q-24 AD7569SQ2 –55(cid:176) C to +125(cid:176) C – 1 LSB Q-24 AD7569BN –40(cid:176) C to +85(cid:176) C – 0.5 LSB N-24 AD7569KN 0(cid:176) C to +70(cid:176) C – 0.5 LSB N-24 AD7569BR –40(cid:176) C to +85(cid:176) C – 0.5 LSB R-24 AD7569BQ –40(cid:176) C to +85(cid:176) C – 0.5 LSB Q-24 AD7569TQ2 –55(cid:176) C to +125(cid:176) C – 1/2 LSB Q-24 AD7569JP 0(cid:176) C to +70(cid:176) C – 1 LSB P-28A AD7569SE2 –55(cid:176) C to +125(cid:176) C – 1 LSB E-28A AD7569KP 0(cid:176) C to +70(cid:176) C – 1/2 LSB P-28A AD7569TE2 –55(cid:176) C to +125(cid:176) C – 1/2 LSB E-28A AD7669AN –40(cid:176) C to +85(cid:176) C – 1 LSB N-28 PLCC AD7669JN 0(cid:176) C to +70(cid:176) C – 1 LSB N-28 AD7669JP 0(cid:176) C to +70(cid:176) C – 1 LSB P-28A AD7669AR –40(cid:176) C to +85(cid:176) C – 1 LSB R-28 AD7669JR 0(cid:176) C to +70(cid:176) C – 1 LSB R-28 NOTES 1E = Leadless Ceramic Chip Carrier; N = Plastic DIP; P = Plastic Leaded Chip Carrier; Q = Cerdip; R = Small Outline SOIC. 2To order MIL-STD-883, Class B processed parts, add /883B to part number. Contact your local sales office for military data sheet. –6– REV. B

AD7569/AD7669 PIN FUNCTION DESCRIPTION (Applies to the AD7569 and AD7669 unless otherwise stated.) Pin Pin Mnemonic Description Mnemonic Description AGND Analog Ground for the DAC(s). Separate CS Chip Select Input (Active Low). The device is DAC ground return paths are provided for the selected when this input is active. DAC(s) and ADC to minimize crosstalk. RD READ Input (Active Low). This input must VOUT Output Voltage. VOUT is the buffered output be active to access data from the part. In the (VOUTA, VOUTB) voltage from the AD7569 DAC. VOUTA and Mode 2 interface, RD going low starts con- VOUTB are the buffered DAC output voltages version. It is used in conjunction with the CS from the AD7669. Four different output volt- input (see Digital Interface Section). age ranges can be achieved (see Table I). ST Start Conversion (Edge triggered). This is VSS Negative Supply Voltage (–5 V for dual sup- used when precise sampling is required. The ply or 0 V for single supply). This pin is also falling edge of ST starts conversion and drives used with the RANGE pin to select the differ- BUSY low. The ST signal is not gated with ent input/output ranges and changes the data CS. format from binary (V = 0 V) to 2s comple- SS BUSY BUSY Status Output (Active Low). When ment (V = –5 V) (see Table I). SS this pin is active, the ADC is performing a RANGE Range Selection Input. This is used with the conversion. The input signal is held prior to VSS input to select the different ranges as per the falling edge of BUSY (see Digital Inter- Table I. The range selected applies to both face Section). the analog input voltage of the ADC and the INT INTERRUPT Output (Active Low). INT go- output voltage from the DAC(s). ing low indicates that the conversion is com- RESET Reset Input (Active Low). This is an asyn- plete. INT goes high on the rising edge of CS chronous system reset that clears the DAC or RD and is also set high by a low pulse on register(s) to all 0s and clears the INT line of RESET (see Digital Interface Section). the ADC (i.e., makes the ADC ready for new A/B (AD7669 DAC Select Input. This input selects which conversion). In unipolar operation, this input Only) DAC register data is written to under control sets the output voltage to 0 V; in bipolar of CS and WR. With this input low, data is operation, it sets the output to negative full written to the DACA register; with this input scale. high, data is written to the DACB register. DB7 Data Bit 7. Most Significant Bit (MSB). CLK A TTL compatible clock signal may be used DB6–DB2 Data Bit 6 to Data Bit 2. to determine the ADC conversion time. Inter- DGND Digital Ground. nal clock operation is achieved by connecting a resistor and capacitor to ground. DB1 Data Bit 1. AGND Analog Ground for the ADC. DB0 Data Bit 0. Least Significant Bit (LSB). ADC V Analog Input. Various input ranges can be se- WR Write Input (Edge triggered). This is used in IN lected (see Table I). conjunction with CS to write data into the AD7569 DAC register. It is used in conjunc- VDD Positive Supply Voltage (+5 V). tion with CS and A/B to write data into the selected DAC register of the AD7669. Data is transferred on the rising edge of WR. Table I. Input/Output Ranges Input/Output DB0–DB7 Range V Voltage Range Data Format SS 0 0 V 0 V to +1.25 V Binary 1 0 V 0 V to +2.5 V Binary 0 –5 V – 1.25 V 2s Complement 1 –5 V – 2.5 V 2s Complement REV. B –7–

AD7569/AD7669—Typical Performance Graphs Noise Spectral Density vs. Frequency Power Supply Rejection Ratio vs. Frequency Positive-Going Settling Time (– 2.5 V Range) Negative-Going Settling Time (– 2.5 V Range) DAC/ADC Full-Scale Temperature Coefficient IMD Plot for ADC –8– REV. B

AD7569/AD7669 CIRCUIT DESCRIPTION supply, a transistor on the output acts as a passive pull-down D/A SECTION with output voltages near 0 V with V = 0 V. This means that SS The AD7569 contains an 8-bit, voltage-mode, D/A converter the sink capability of the amplifier is reduced as the output volt- that uses eight equally weighted current sources switched into age nears 0 V in single supply. In dual supply operation the full an R-2R ladder network to give a direct but unbuffered 0 V to sink capability of 1.25 mA is maintained over the entire output +1.25 V output range. The AD7669 is similar, but contains two voltage range. D/A converters. The current sources are fabricated using PNP For all other parameters, the single and dual supply perfor- transistors. These transistors allow current sources that are mances of the amplifier are essentially identical. The output driven from positive voltage logic and give a zero-based output noise from the amplifier, with full scale on the DAC, is 200 m V range. The output voltage from the voltage switching R-2R lad- peak-to-peak. The spot noise at 1 kHz is 35 nV/(cid:214) Hz with all 0s der network has the same positive polarity as the reference; on the DAC. A noise spectral density versus frequency plot for therefore, the D/A converter can be operated from a single the amplifier is shown in the typical performance graphs. power supply rail. VOLTAGE REFERENCE The PNP current sources are generated using the on-chip The AD7569/AD7669 contains an on-chip bandgap reference bandgap reference and a control amplifier. The current sources that provides a low noise, temperature compensated reference are switched to either the ladder or AGND by high speed DAC voltage for both the DAC and the ADC. The reference is p-channel switches. These high-speed switches ensure a fast set- trimmed for absolute accuracy and temperature coefficient. The tling time for the output voltage of the DAC. The R-2R ladder bandgap reference is generated with respect to V . It is buff- network of the DAC consists of highly stable, thin-film resistors. DD ered by a separate control amplifier for both the DAC and the A simplified circuit diagram for the D/A converter section is ADC reference. This can be seen in the DAC ladder network shown in Figure 3. An identical D/A converter is used as part of configuration in Figure 3. the A/D converter, which is discussed later. DIGITAL SECTION The data pins on the AD7569/AD7669 provide a connection between the external bus and DAC data inputs and ADC data outputs. The threshold levels of all digital inputs and outputs are compatible with either TTL or 5 V CMOS levels. Internal input protection of all digital pins is achieved by on-chip distrib- uted diodes. The data format is straight binary when the part is used in single supply (V = 0 V). However, when a V of –5 V is applied, the SS SS data format becomes twos complement. This data format ap- plies to the digital inputs of the DAC and the digital outputs of the ADC. ADC SECTION The analog-to-digital converter on the AD7569/AD7669 uses Figure 3.DAC Simplified Circuit Diagram the successive approximation technique to achieve a fast conver- sion time of 2 m s and provides an 8-bit parallel digital output. OP AMP SECTION The reference for the ADC is provided by the on-chip bandgap The output from the D/A converter is buffered by a high speed, reference. noninverting op amp. This op amp is capable of developing – 2.5 V across a 2 kW and 100 pF load to AGND . The am- Conversion start is controlled by ST or by CS and RD. Once a DAC plifier can be operated from a single +5 V supply to give two conversion has been started, another conversion start should not unipolar output ranges, or from dual supplies (– 5 V) to allow be attempted until the conversion in progress is completed. two bipolar output ranges. Exercising the RESET input does not affect conversion; the RESET input resets the INT line high, which is useful in inter- The feedback path of the amplifier contains a gain/offset net- rupt driven systems where a READ has not been performed at work that provides four voltage ranges at the output of the op the end of the previous conversion. The INT line does not have amp. The output voltage range is determined by the RANGE to be cleared at the end of conversion. The ADC will continue and V inputs. (See Table I in the Pin Function Description SS to convert correctly, but the function of the INT line will be section.) The four possible output ranges are: 0 V to +1.25 V, 0 V to +2.5 V, – 1.25 V and – 2.5 V. It should be noted that affected. whichever range is selected for the output amplifier also applies Figure 4 shows the operating waveforms for a conversion cycle. to the input voltage range of the A/D converter. The analog input voltage, V , is held 50 ns typical after the fall- IN ing edge of ST or (CS & RD). The MSB decision is made ap- The output amplifier settles to within 1/2 LSB of its final value proximately 50 ns after the second falling edge of the input in typically less than 500 ns. Operating the part from single or CLK following a conversion start. If t in Figure 4 is greater dual supplies has no effect on the positive-going settling time. 1 than 50 ns, then the falling edge of the input CLK will be seen However, the negative-going output settling time to voltages as the first falling clock edge. If t is less than 50 ns, the first fall- near 0 V in single supply will be slightly longer than the settling 1 ing clock edge of the conversion will not occur until one clock time to negative full scale for dual supply operation. Addition- cycle later. The succeeding bit decisions are made approxi- ally, to ensure that the output voltage can go to 0 V in single mately 50 ns after a CLK edge until conversion is complete. REV. B –9–

AD7569/AD7669 At the end of conversion, the SAR contents are transferred to INTERNAL CLOCK the output latch, and the SAR is reset in readiness for a new Clock pulses are generated by the action of an internal current conversion. A single conversion lasts for 8 input clock cycles. source charging the external capacitor (C ) and this external CLK capacitor discharging through the external resistor (R ). CLK When a conversion is complete, this internal clock stops operat- ing and the CLK pin goes to the DGND potential. Connections for R and C are shown in the operating diagram of Fig- CLK CLK ure 21. The nominal conversion time versus temperature for the recommended R and C combination is shown in Figure CLK CLK 6. The internal clock provides a convenient clock source for the AD7569/AD7669. Due to process variations, the actual operat- ing frequency for this R /C combination can vary from CLK CLK device to device by up to – 25%. Figure 4.Operating Waveforms Using External Clock ANALOG INPUT The analog input of the AD7569/AD7669 feeds into an on-chip track-and-hold amplifier. To accommodate different full-scale ranges, the analog input signal is conditioned by a gain/offset network that conditions all input ranges so the internal ADC al- ways works with a 0 V to +1.25 V signal. As a result, the input current on the V input varies with the input range selected as IN shown in Figure 5. Figure 6.Conversion Time vs. Temperature for Internal Clock Operation DIGITAL INTERFACE Figure 5.Equivalent V Circuit IN DAC Timing and Control—AD7569 TRACK-AND-HOLD Table II shows the truth table for DAC operation for the The track-and-hold (T/H) amplifier on the analog input of the AD7569. The part contains an 8-bit DAC register, which is AD7569/AD7669 allows the ADC to accurately convert an in- loaded from the data bus under control of CS and WR. The put sine wave of 2.5 V peak-to-peak amplitude up to a fre- data contained in the DAC register determines the analog out- quency of 200 kHz, the Nyquist frequency of the ADC when put from the DAC. The WR input is an edge-triggered input, operated at its maximum throughput rate of 400 kHz. This and data is transferred into the DAC register on the rising edge maximum rate of conversion includes conversion time and time of WR. Holding CS and WR low does not make the DAC regis- between conversions. Because the input bandwidth of the T/H ter transparent. amplifier is much larger than 200 kHz, the input signal should be band-limited to avoid converting high-frequency noise Table II. AD7569 DAC Truth Table components. CS WR RESET DAC Function The operation of this T/H amplifier is essentially transparent to the user. The T/H amplifier goes from its tracking mode to its H H H DAC Register Unaffected hold mode at the start of conversion. This occurs when the L L H DAC Register Unaffected ADC receives a conversion start command from either ST or L g H DAC Register Updated CS & RD. At the end of conversion (BUSY going high), the g L H DAC Register Updated T/H reverts back to tracking the input signal. X X L DAC Register Loaded with All Zeros EXTERNAL CLOCK L = Low State, H = High State, X = Don’t Care The AD7569/AD7669 ADC can be used with its on-chip clock or with an externally applied clock. When using an external The contents of the DAC register are reset to all 0s by an active clock, the CLK input of the AD7569/AD7669 may be driven low pulse on the RESET line, and for the unipolar output ranges, directly from 74HC, 4000B series buffers (such as 4049) or the output remains at 0 V after RESET returns high. For the bi- from TTL buffers. When conversion is complete, the internal polar output ranges, a low pulse on RESET causes the output to clock is disabled. The external clock can continue to run be- go to negative full scale. In unipolar applications, the RESET line tween conversions without being disabled. The mark/space ratio can be used to ensure power-up to 0 V on the AD7569 DAC out- of the external clock can vary from 70/30 to 30/70. put and is also useful when used as a zero override in system cali- bration cycles. If the RESET input is connected to the system –10– REV. B

AD7569/AD7669 RESET line, the DAC output resets to 0 V when the entire The contents of the DAC registers are reset to all 0s by an active system is reset. Figure 7 shows the input control logic for the low pulse on the RESET line, and for the unipolar output AD7569 DAC; the write cycle timing diagram is shown in ranges, the outputs remain at 0 V after RESET returns high. Figure 8. For the bipolar output ranges, a low pulse on RESET causes the outputs to go to negative full scale. In unipolar applications, the RESET line can be used to ensure power-up to 0 V on the AD7669 DAC outputs and is also useful when used as a zero override in system calibration cycles. If the RESET input is con- nected to the system RESET line, then the DAC outputs reset to 0 V when the entire system is reset. Figure 9 shows the DAC input control logic for the AD7669, and the write cycle timing diagram is shown in Figure 8. Figure 7.AD7569 DAC Input Control Logic Figure 9.AD7669 DAC Control Logic ADC Timing and Control The ADC on the AD7569/AD7669 is capable of two basic oper- ating modes. In the first mode, the ST line is used to start con- version and drive the track-and-hold into hold mode. At the end of conversion, the track-and-hold returns to its tracking mode. The second mode is achieved by hard-wiring the ST line high. In this case, CS and RD start conversion, and the microproces- Figure 8.AD7569/AD7669 Write Cycle Timing Diagram sor is driven into a WAIT state for the duration of conversion by DAC Timing and Control—AD7669 BUSY. Table III shows the truth table for the dual DAC operation of the AD7669. The part contains two 8-bit DAC registers that are loaded from the data bus under the control of CS, A/B and WR. Address line A/B selects which DAC register the data is loaded to. The data contained in the DAC registers determines the analog output from the respective DACs. The WR input is an edge-triggered input, and data is transferred into the selected DAC register on the rising edge of WR. Holding CS and WR low does not make the selected DAC register transparent. The A/B input should not be changed while CS and WR are low. Table III. AD7669 DAC Truth Table CS WR A/B RESET DAC Function H H X H DAC Registers Unaffected L g L H DACA Register Updated g L L H DACA Register Updated L g H H DACB Register Updated g L H H DACB Register Updated X X X L DAC Registers Loaded with All Zeros Figure 10.ADC Mode 1 Interface Timing L = Low State, H = High State, X = Don’t Care REV. B –11–

AD7569/AD7669 MODE 1 INTERFACE MODE 2 INTERFACE The timing diagram for the first mode is shown in Figure 10. It The second interface mode is intended for use with micropro- can be used in digital signal processing and other applications cessors, which can be forced into a WAIT state for at least 2 m s. where precise sampling in time is required. In these applica- The ST line of the AD7569/AD7669 must be hardwired high to tions, it is important that the signal sampling occurs at exactly achieve this mode. The microprocessor starts a conversion and equal intervals to minimize errors due to sampling uncertainty is halted until the result of the conversion is read from the con- or jitter. In these cases, the ST line is driven by a timer or some verter. Conversion is initiated by executing a memory READ to precise clock source. the AD7569/AD7669 address, bringing CS and RD low. BUSY subsequently goes low (forcing the microprocessor READY or The falling edge of the ST pulse starts conversion and drives the WAIT input low), placing the microprocessor into a WAIT AD7569/AD7669 track-and-hold amplifier into its hold mode. state. The input signal is held on the falling edge of RD (assum- BUSY stays low for the duration of conversion and returns high ing CS is already low or is coincident with RD). When the con- at the end of conversion and the track-and hold amplifier reverts version is complete (BUSY goes high), the processor completes to its tracking mode on this rising edge of BUSY. The INT line the memory READ and acquires the newly converted data. can be used to interrupt the microprocessor. A READ to the While conversion is in progress, the ADC places old data (from AD7569/AD7669 address accesses the data, and the INT line is the previous conversion) on the data bus. The timing diagram reset on the rising edge of CS or RD. Alternatively, the INT can for this interface is shown in Figure 12. be used to trigger a pulse that drives the CS and RD and places the data into a FIFO or buffer memory. The microprocessor can then read a batch of data from the FIFO or buffer memory at some convenient time. The ST input should not be high when RD is brought low; otherwise, the part will not operate correctly in this mode. It is important, especially in systems where the conversion start (ST pulse) is asynchronous to the microprocessor, that a READ does not occur during a conversion. Trying to read data from the device during a conversion can cause errors to the conver- sion in progress. Also, pulsing the ST line a second time before conversion ends should be avoided since it too can cause errors in the conversion result. In applications where precise sampling is not critical, the ST pulse can be generated from a micropro- cessor WR or RD line gated with a decoded address (different from AD7569/AD7669 CS address). Figure 12.ADC Mode 2 Interface Timing The major advantage of this interface is that it allows the micro- processor to start conversion, WAIT, and then READ data with a single READ instruction. The user does not have to worry about servicing interrupts or ensuring that software delays are long enough to avoid reading during conversion. The fast con- version time of the ADC ensures that for many microprocessors, the processor is not placed in a WAIT state for an excessive amount of time. DIGITAL SIGNAL PROCESSING APPLICATIONS In Digital Signal Processing (DSP) application areas such as voice recognition, echo cancellation and adaptive filtering, the Figure 11.Multichannel Inputs dynamic characteristics (SNR, Harmonic Distortion, Intermod- ulation Distortion) of both the ADC and DAC are critical. The This interface mode is also useful in applications where a num- AD7569/AD7669 is specified dynamically as well as with stan- ber of input channels are required to be converted by the ADC. dard dc specifications. Because the track/hold amplifier has a Figure 11 shows the circuit configuration for such an applica- wide bandwidth, an antialiasing filter should be placed on the tion. The signal that drives the ST input of the AD7569/ V input to avoid aliasing of high-frequency noise back into the AD7669 is also used to drive the ENABLE input of the multi- IN band of interest. plexer. The multiplexer is enabled on the rising edge of the ST pulse while the input signal is held on the falling edge; therefore, The dynamic performance of the ADC is evaluated by applying a the signal must have settled to within 8 bits over the duration of sine-wave signal of very low distortion to the V input, which is IN this ST pulse. The settling time, including t (ENABLE) of sampled at a 409.6 kHz sampling rate. A Fast Fourier Transform ON the multiplexer plus the T/H acquisition time (typically 200 ns), (FFT) plot or Histogram plot is then generated from which SNR, thus determines the width of the ST pulse. This is suited to ap- harmonic distortion and dynamic differential nonlinearity data plications where a number of input channels needs to be succes- can be obtained. For the DAC, the codes for an ideal sine wave sively sampled or scanned. are stored in PROM and loaded down to the DAC. The output spectrum is analyzed, using a spectrum analyzer to evaluate SNR –12– REV. B

AD7569/AD7669 and harmonic distortion performance. Similarly, for inter- modulation distortion, an input (either to V or DAC code) IN consisting of pure sine waves at two frequencies is applied to the AD7569/AD7669. Figure 15.DAC Output Spectrum HISTOGRAM PLOT When a sine wave of specified frequency is applied to the V in- IN put of the AD7569/AD7669 and several thousand samples are taken, it is possible to plot a histogram showing the frequency of occurrence of each of the 256 ADC codes. If a particular step is wider than the ideal 1 LSB width, the code associated with that step will accumulate more counts than for the code for an ideal step. Likewise, a step narrower than ideal width will have fewer Figure 13.ADC FFT Plot counts. Missing codes are easily seen because a missing code means zero counts for a particular code. The absence of large Figure 13 shows a 2048 point FFT plot of the ADC with an in- spikes in the plot indicates small differential nonlinearity. put signal of 130 kHz. The SNR is 48.4 dB. It can be seen that most of the harmonics are buried in the noise floor. It should be Figure 16 shows a histogram plot for the ADC indicating very noted that the harmonics are taken into account when calculat- small differential nonlinearity and no missing codes for an input ing the SNR. The relationship between SNR and resolution (N) frequency of 204 kHz. For a sine-wave input, a perfect ADC is expressed by the following equation: would produce a cusp probability density function described by the equation SNR = (6.02N + 1.76) dB (cid:49) (cid:112)(cid:40)(cid:86)(cid:41)= This is for an ideal part with no differential or integral linearity p (cid:40)(cid:65)(cid:50)- (cid:86)(cid:50)(cid:41)(cid:49)(cid:47)(cid:50) errors. These errors will cause a degradation in SNR. By work- ing backward from the above equation, it is possible to get a where A is the peak amplitude of the sine wave and p(V) the measure of ADC performance expressed in effective number of probability of occurrence at a voltage V. bits (N). This effective number of bits is plotted versus fre- The histogram plot of Figure 16 corresponds very well with this quency in Figure 14. The effective number of bits typically falls cusp shape. between 7.7 and 7.8, corresponding to SNR figures of 48.1 dB Further typical plots of the performance of the AD7569/AD7669 and 48.7 dB. are shown in the Typical Performance Graphs section of the data Figure 15 shows a spectrum analyzer plot of the output spec- sheet. trum from the DAC with an ideal sine-wave table loaded to the data inputs of the DAC. In this case, the SNR is 46 dB. Figure 14.Effective Number of Bits vs. Frequency Figure 16.ADC Histogram Plot REV. B –13–

AD7569/AD7669 INTERFACING THE AD7569/AD7669 AD7569/AD7669—ADSP-2100 INTERFACE AD7569/AD7669—Z80 INTERFACE Figure 19 shows a typical interface to the DSP processor, the Figure 17 shows a typical interface to the Z80 microprocessor. ADSP-2100. The ADC is in the Mode 2 interface mode, which The ADC is configured for operation in the Mode 1 interface means that the ADSP-2100 is halted during conversion. This is mode. A precise timer or clock source starts conversion in appli- achieved using the decoded address output. This is gated with cations requiring equidistant sampling intervals. The scheme DMWR to ensure that it halts the processor for READ instruc- used, whereby INT of the AD7569/AD7669 generates an inter- tions only. INT going low at the end of conversion releases the rupt on the Z80, is limited in that it does not allow the ADC to processor and allows it to finish off the READ instruction. be sampled at the maximum rate. This is because the time be- tween samples has to be long enough to allow the Z80 to service its interrupt and read data from the ADC. To overcome this, some buffer memory or FIFO could be placed between the AD7569/AD7669 and the Z80. Writing data to the relevant AD7569/AD7669 DAC simply consists of a <LD (nn), A> in- struction where nn is the decoded address for that DAC. Read- ing data from the ADC, after an INT has been received, consists of a < LDA, (nn)> instruction. Figure 19.AD7569/AD7669 to ADSP-2100 Interface Because the instruction cycle of the ADSP-2100 is so fast (125ns cycle), the DMWR pulse also has to be stretched also for write cycles. This is achieved using the 74121, which gener- ates a pulse that is fed back to DMACK. The duration of this pulse determines how long the ADSP-2100 write cycle is stretched. The buffers driving the DMACK line must have Figure 17.AD7569/AD7669 to Z80 Interface open-collector outputs. Writing data to the relevant AD7569/ AD7569/AD7669—68008 INTERFACE AD7669 DAC is achieved using a single instruction, <DM A typical interface to the 68008 is shown in Figure 18. In this (addr) = MRO>, where addr is the decoded address of that case, the ADC is configured in the Mode 2 interface mode. This DAC, and MRO contains the data to be loaded to the DAC reg- means that the one read instruction starts conversion and reads ister. Data is read from the ADC also, using a single instruction the data. The read cycle is stretched out over the entire conver- <MRO = DM (addr)>, where the conversion result is placed in sion period by taking the INT line back into the DTACK input the MRO data register. of the 68008. The additional gates are required so the 68008 AD7569/AD7669—IBM PC* INTERFACE receives a DTACK when the processor is writing data to the The AD7569/AD7669 is ideal for implementing an analog in- AD7569/AD7669. In this case, there are no wait states intro- put/output port for the IBM PC. Figure 20 shows an interface duced into the write cycle. Writing data to the relevant AD7569/ that realizes this function. The ADC is configured in the Mode AD7669 DAC consists of a <MOVE.B Dn, addr> where Dn is 1 interface mode, and conversions are initiated using a precise the data register, which contains the data to be loaded to that clock source for equidistant sampling intervals. At the end of DAC, and addr is the decoded address for the DAC. Data is conversion, the INT line goes low, and the 74121 generates read from the ADC using a <MOVE.B addr,Dn> with the con- version result placed in register Dn. Figure 20.AD7569/AD7669 to IBM PC Interface Figure 18.AD7569/AD7669 to 68008 Interface *IBM PC is a trademark of International Business Machines Corp. –14– REV. B

AD7569/AD7669 an RD pulse for the AD7569/AD7669. This RD pulse accesses UNIPOLAR (0 V to +2.5 V) CONFIGURATION data from the ADC and places the conversion result into a regis- The 0 V to +2.5 V output voltage range is achieved by tying V SS ter on the 74646. The rising edge of this pulse generates an in- to AGND (= 0 V) and the RANGE input to V . The table DAC DD terrupt request to the processor. The conversion result is read for output voltage versus digital code is as in Table IV with from the 74646 register by performing an I/O read to the 2.V replacing V . Note that for this range REF REF decoded address of the 74646. Writing data to the relevant AD7569/AD7669 DAC involves an I/O write to the 74646, (cid:49)(cid:32)(cid:76)(cid:83)(cid:66)=(cid:50)(cid:46)(cid:86) (cid:40)(cid:50)- (cid:56)(cid:41)=(cid:86) (cid:49) which transfers the data to the data inputs of the AD7569/ (cid:82)(cid:69)(cid:70) (cid:82)(cid:69)(cid:70)(cid:49)(cid:50)(cid:56) AD7669. Data is latched into the selected DAC register on the BIPOLAR (–1.25 V to +1.25 V) CONFIGURATION rising edge of IOW. The first of the bipolar configurations is achieved by tying the APPLYING THE AD7569/AD7669 DAC RANGE input to AGND (= 0 V) and V to –5 V. The V DAC SS SS An internal gain/offset network on the AD7569/AD7669 allows voltage level at which the AD7569/AD7669 changes to bipolar several output voltage ranges. The part can produce unipolar operation is approximately –1 V. When the part is configured output ranges of 0 V to +1.25 V or 0 V to +2.5 V and bipolar for bipolar outputs, the input coding becomes twos comple- output ranges of –1.25 V to +1.25 V or –2.5 V to +2.5 V. Con- ment. The table for output voltage versus the digital code in the nections for these various output ranges are outlined below. DAC register is shown in Table V. Note as with the unipolar UNIPOLAR (0 V to +1.25 V) CONFIGURATION configuration, a digital input code of all 0s produces an output The first of the configurations provides an output voltage range of 0 V. It should be noted, however, that a low pulse on the of 0 V to +1.25 V. This is achieved by tying the VSS and RESET line for the bipolar ranges sets the output voltage to RANGE inputs to AGNDDAC(= 0 V). Figure 21 shows the con- negative full scale. figuration of the AD7569 to achieve this output range. A similar Table V. Bipolar (–1.25 V to +1.25 V) Code Table configuration of the AD7669 gives the same output range. The table for output voltage versus the digital code in the DAC regis- DAC Register Contents ter is shown in Table IV. MSB LSB Analog Output, V OUT (cid:230) (cid:49)(cid:50)(cid:55)(cid:246) 0111 1111 +V (cid:231) (cid:247) REF Ł (cid:49)(cid:50)(cid:56)ł (cid:230) (cid:49) (cid:246) 0000 0001 +V (cid:231) (cid:247) REF Ł (cid:49)(cid:50)(cid:56)ł 0000 0000 0 V (cid:230) (cid:49) (cid:246) 1111 1111 –V (cid:231) (cid:247) REF Ł (cid:49)(cid:50)(cid:56)ł (cid:230) (cid:49)(cid:50)(cid:55)(cid:246) 1000 0001 –V (cid:231) (cid:247) REF Ł (cid:49)(cid:50)(cid:56)ł Figure 21.AD7569 Unipolar (0 V to +1.25 V) Operation (cid:230) (cid:49)(cid:50)(cid:56)(cid:246) 1000 0000 –V (cid:231) (cid:247) = –V Table IV. Unipolar (0 V to +1.25 V) Code Table REF Ł (cid:49)(cid:50)(cid:56)ł REF NOTE: 1 LSB = (V )(2–7) = V (1/128) DAC Register Contents REF REF MSB LSB Analog Output, VOUT BIPOLAR (–2.5 V to +2.5 V) CONFIGURATION The –2.5 V to +2.5 V bipolar output range is achieved by tying (cid:230) (cid:50)(cid:53)(cid:53)(cid:246) the RANGE input to V and the V input to –5 V. Once 1111 1111 +V (cid:231) (cid:247) DD SS REF Ł (cid:50)(cid:53)(cid:54)ł again, the input coding is 2s complement. The table for output voltage versus digital code is as in Table V with 2.V replacing (cid:230) (cid:49)(cid:50)(cid:57)(cid:246) REF 1000 0001 +VREF Ł(cid:231) (cid:50)(cid:53)(cid:54)ł(cid:247) VREF. Note that for this range 1000 0000 +V (cid:230)(cid:231) (cid:49)(cid:50)(cid:56)(cid:246)(cid:247) = +V /2 (cid:49) (cid:32)(cid:76)(cid:83)(cid:66)=(cid:52)(cid:46)(cid:86)(cid:82)(cid:69)(cid:70)(cid:40)(cid:50)- (cid:56)(cid:41)=(cid:86)(cid:82)(cid:69)(cid:70) (cid:54)(cid:49)(cid:52) REF Ł (cid:50)(cid:53)(cid:54)ł REF (cid:230) (cid:49)(cid:50)(cid:55)(cid:246) 0111 1111 +V (cid:231) (cid:247) REF Ł (cid:50)(cid:53)(cid:54)ł (cid:230) (cid:49) (cid:246) 0000 0001 +V (cid:231) (cid:247) REF Ł (cid:50)(cid:53)(cid:54)ł 0000 0000 0 V NOTE: 1 LSB = (V ) (2–8) = V (1/256); V = +1.25 V Nominal REF REF REF REV. B –15–

AD7569/AD7669 APPLYING THE AD7569/AD7669 ADC The analog input on the AD7569/AD7669 accepts the same four input ranges as the output ranges on the DAC. Whatever output range is selected for the DAC also applies to the input range of the ADC. Although separate AGNDs exist for both the DAC and ADC to minimize crosstalk, writing data to the DAC while the ADC is performing a conversion may result in an incorrect conversion from the ADC due to an interaction of currents between the DAC and ADC. Therefore, to ensure correct operation of the ADC, the DAC register should not be updated while the ADC is converting. UNIPOLAR OPERATION The circuit of Figure 21 shows the AD7569 configured for both an input and output range of 0 V to +1.25 V (the AD7669 con- figuration is similar). The nominal transfer characteristic for this range is shown in Figure 22. The output code is Natural Binary Figure 23.Nominal Transfer Characteristic for Bipolar with 1 LSB = (1.25/256)V = 4.88 mV. (–1.25 V to +1.25 V) Operation As before, to achieve the unipolar 0 V to +2.5 V input range, typical example is a digital filter where an ac analog signal is V is connected to 0 V, and the RANGE input is tied to a logic quantized by the ADC, digitally processed and recreated using SS high. The nominal transfer characteristic is as in Figure 22 but, the DAC. In these types of applications, the offset error can be in this case, 1 LSB = (2.5/256)V = 9.76 mV. eliminated by ac coupling the recreated signal. Full-scale error effect is linear and does not cause problems as long as the input signal is within the full dynamic range of the ADC. An impor- tant parameter in DSP applications is Differential Nonlinearity, and this is not affected by either offset or full-scale error. In applications where absolute accuracy is important ADC off- set and full-scale error can be adjusted to zero. Figure 24 shows the additional components required for offset and full-scale er- ror adjustment. Offset error must be adjusted before full-scale error. Zero offset is achieved by adjusting the offset of the op amp driving V (i.e., A1 in Figure 23). In unipolar applica- IN tions, for zero offset error, apply 1/2 LSB at the analog input and adjust the op amp offset voltage until the ADC output code flickers between 0000 0000 and 0000 0001. For zero full-scale error, apply an analog input of FS – 3/2 LSBs and adjust R1 un- til the ADC output code flickers between 1111 1110 and 1111 1111. In bipolar applications, to adjust for bipolar zero offset, apply –1/2 LSB at the analog input and adjust the op amp offset volt- Figure 22.Nominal Transfer Characteristic for Unipolar age until the output code flickers between 1111 1111 and 0000 (0 V to +1.25 V) Operation 0000. For zero full-scale error, apply +FS/2 – 3/2 LSB at the analog input and adjust R1 until the ADC output code flickers BIPOLAR OPERATION between 0111 1110 and 0111 1111. The analog input of the AD7569/AD7669 ADC is configured for bipolar inputs when V = –5 V. The output code provided SS by the part is twos complement. Figure 23 shows the transfer function for bipolar (–1.25 V to +1.25 V) operation. The LSB size for this range is (2.5/256)V = 9.76 mV. The transfer function for the –2.5 V to +2.5 V range is identical to that of Figure 23, but now FS = 5 V and the LSB size is (5/256)V = 19.5 mV. ADC OFFSET AND FULL-SCALE ERROR ADJUSTMENT In most Digital Signal Processing (DSP) applications, offset and full-scale error have little or no effect on system performance. A Figure 24.ADC Error Adjust Circuit –16– REV. B

AD7569/AD7669 Figure 25.Peak-Reading A/D Converter PEAK DETECTION—AD7569 head (or motor) is monitored. The closed-loop system allows an The circuit of Figure 25 shows a peak-reading A/D converter, error between the desired position and the actual position to be which is useful in such applications as monitoring flow rates, monitored and corrected. The correction is achieved by adjust- temperature, pressure, etc. The circuit ensures that a peak will ing the ratio of the phase currents in the motor windings until not be missed while at the same time does not require the mi- the required head position is reached. croprocessor to frequently monitor the data. The peak value is The AD7669 is ideally suited for the closed-loop microstepping stored in the A/D converter and can be read at any time. technique with its on-chip dual DACs for positioning the disk The gain on the AD524 is adjusted to yield a 0 V to +2.5 V out- drive head, and onboard ADC for monitoring the position of the put. When the input signal exceeds the current stored value, the head. A generalized circuit for a closed-loop microstepping sys- output of the TL311 goes low, triggering the Q output of the tem is shown in Figure 26. The DAC waveforms are shown in 74121. This low-going pulse starts a conversion on the AD7569 Figure 27, along with the direction information for clockwise ro- ADC, and at the end of conversion latches the result into the tation supplied by the controller. DAC. This pulse must be at least 120 ns greater than the con- version time of the ADC. The Q output is used to drive the strobe input of the TL311, resetting the TL311 output high in readiness for another conversion. The additional gates on the RD and WR inputs are to allow the data to be read by the microprocessor while at the same time ensuring that the DAC is not updated when the microprocessor reads the data. It may be necessary to monitor the AD7569 BUSY line to ensure that a processor READ is not attempted while the AD7569 is in the middle of a conversion. The READ pulse width from the processor must be less than 1 m s to ensure correct data is read from the ADC. A low-going pulse on the RESET line resets the DAC output to 0 V and starts a new “peak- detection” period. This RESET pulse must also be less than 1 m s. DISK DRIVE APPLICATION—AD7669 Closed-Loop Microstepping Microstepping is a popular technique in low density disk drives Figure 26. Typical Closed-Loop Microstepping Circuit with (both floppy and hard disk) that allows higher positional resolu- the AD7669 tion of the disk drive head over that obtainable from a full- step The AD7669 is used in the unipolar 0 V to +2.5 V configura- driven stepper motor. Typically, a two-phase stepper motor has tion. This allows the circuit of Figure 26 to be completely uni- its phase currents driven with a sine-cosine relationship. These polar (+5 V, +12 V supplies); no negative power supplies are cosinusoidal signals are generated by two DACs driven with the required. The power output stage is a dual H-Bridge device appropriate data. The resolution of the DACs determines the such as the UDN-2998W from Sprague Electric. The phase number of microsteps into which each full step can be divided. currents in both windings are detected by means of the small For example, with a 1.8(cid:176) full-step motor and a 4-bit DAC, a value sense resistors, R A and R B, in series with the windings. microstep size of 0.11(cid:176) (1.8(cid:176) /(2n)) is obtainable. S S The voltage developed across these resistors is amplified and The microstepping technique improves the positioning resolu- compared with the respective DAC output voltage. The com- tion possible in any control application; however, the positional parators in turn chop the phase winding current. The ADC accuracy can be significantly worse than that offered by the completes the feedback path by converting information from a original full-step accuracy specification due to load torque effects. suitable transducer for analysis by the controller. To ensure that the increased resolution is usable, it is necessary to use a closed-loop system where the position of the disk drive REV. B –17–

AD7569/AD7669 On initial start-up, the output voltage, V , will be invalid until O the length of the delay is reached (i.e., until the counter is re- set). From this point forward, the delayed data is read from the 6116 and loaded to the DAC before the newly converted data is written into the same memory location. The input clock to the system can be a square wave of maximum input frequency 200 kHz (assuming 2 m s conversion time for the ADC). The mark/space ratio of the input clock can be varied to maximize the sampling frequency if required. The clock low time has to be equal to the conversion time and access time of the ADC plus the setup time required for the 6116. The clock high time has only to be equal to the setup time for the DAC plus the delay time through the counter and the access time of the 6116. The amount of memory used, as well as the sampling frequency, determines the maximum possible delay. Using the HCT4040, Figure 27.Typical DAC Output Voltages for Microstepping and the 6116 with an input clock frequency of 200 kHz, the and Direction Signals for Clockwise Rotation with the maximum delay is 5 ms on a maximum input frequency of UDN-2998W 100 kHz. Using 64K memory, with an 8 kHz input clock fre- quency, the maximum delay is 8 seconds on a maximum input ANALOG DELAY LINE—AD7569 frequency of 4 kHz. In many applications, especially in audio systems, it is necessary to provide a delay on the input signal. The circuit of Figure 28 TRANSIENT RECORDER—AD7569 shows how a simple analog delay line can be implemented, The scheme just outlined can also form the basis for a transient based on the AD7569. The input signal is sampled using the recorder. In this case, transients on the input signal are con- AD7569 ADC, and converted data is loaded into the 6116 (2K verted and stored in memory. The transient can then be recalled (cid:51) 8 static ram). The inverted input clock drives a counter that from memory at a later time, and the transient waveform can be selects the address for the 6116. The delay is selected by choos- recreated using the AD7569 DAC. ing one of the output lines of the HCT4040 counter to reset the coun-ter. This can be done using a simple switch in a manual INFINITE SAMPLE-AND-HOLD—AD7569 system or by a multiplexer in a programmable delay application. The AD7569 is ideal for implementing a single-chip infinite Data is written to the DAC using the inverted input clock signal. sample-and-hold function. Basically, the ADC samples and con- verts the input signal into an 8-bit digital word. The 8 bits of data are then loaded to the DAC and the sampled value is re- stored to analog form. The sampled value is held until the DAC register is updated. The full-scale matching between the ADC and the DAC on the AD7569 ensures a typical error of less than 1% between the analog input voltage and the “held” output voltage. Figure 29 shows the connections required on the AD7569 to achieve this infinite sample-and-hold function. Figure 29.Infinite Sample-and-Hold Figure 28.Analog Delay Line –18– REV. B

AD7569/AD7669 panel meter module that converts the signal for digital readout. TARE FUNCTION FOR WEIGH SCALE—AD7569 The infinite sample-and-hold just outlined can also form the ba- The input signal to the panel meter is also applied to the analog sis of a circuit to provide a tare function for a weigh scale sys- input of the AD7569 for the tare function. When the tare switch tem. Figure 30 shows a circuit for a weigh scale system. It (S1) is closed, a tare cycle commences and VIN is sampled and incorporates a tare function using a simple circuit based on the held infinitely at VOUT until the next tare cycle. VOUT drives the AD7569. inverting input of the differential amplifier and forces its output to zero. Thus, the tare function is used to give a readout of zero The AD587, along with the 2N6285, provides a buffered +10 V for any undesired weight, such as a box, when only the item reference to supply the low impedance load cell transducer. The placed in it is to be weighed. The tare function can also be used load cell output is amplified by the AD624 precision instrumen- in calibrating the system, to cancel out offset errors due to the tation amplifier with gain adjustment provided by R1. The out- load cell, AD624 and differential amplifier. put of the AD624 is applied to the noninverting input of a unity gain differential summing amplifier that uses the AD707, a high The AD7569 offers many advantages in the system outlined, precision op amp with low drift. The AD707 feeds a 3 1/2 digit such as: simple, low cost circuit—no need for microprocessor, software, etc.—and low power consumption. Figure 30.Weigh Scale System with Tare Function REV. B –19–

AD7569/AD7669 OUTLINE DIMENSIONS Dimensions shown in inches and (mm). 24-Pin Plastic (N-24) 24-Pin Cerdip (Q-24) 8 8 8/ – 0 1 – 4 1 2 1 C 28-Terminal Leadless Ceramic Chip Carrier 28-Terminal Plastic Leaded Chip Carrier (E-28A) (P-28A) 28-Pin Plastic DIP (N-28) 28-Lead Small Outline (SO) (R-28) A. S. U. N D I E T N RI P –20– REV. B